Datasheet LT1169 (Analog Devices) - 8

LT1169
O U U W U APPLICATI S I FOR ATIO
100 CURRENT NOISE = √2qI the total noise. This means the LT1169 is superior to most B 80 dual JFET op amps. Only the lowest noise bipolar op amps 60 have the advantage at low source resistances. As the 40 OP215 source resistance increases from 5k to 50k, the LT1169 20 LT1169 will match the best bipolar op amps for noise perfor- 0 mance, since the thermal noise of the transducer (4kTR) –20 AD822 begins to dominate the total noise. A further increase in –40 INPUT BIAS CURRENT (pA) source resistance, above 50k, is where the op amp’s –60 current noise component (2qI –80 BR2) will eventually domi- –100 nate the total noise. At these high source resistances, the –15 –10 –5 0 5 10 15 LT1169 will out perform the lowest noise bipolar op amps COMMON MODE RANGE (V) due to the inherently low current noise of FET input op LT1169 • F01 amps. Clearly, the LT1169 will extend the range of high
Figure 1. Comparison of LT1169, OP215, and AD822 Input Bias Current vs Common Mode Range
impedance transducers that can be used for high signal- to-noise ratios. This makes the LT1169 the best choice for
Amplifying Signals from High Impedance Transducers
high impedance, capacitive transducers. The low voltage and current noise offered by the LT1169
Optimization Techniques for Charge Amplifiers
makes it useful in a wide range of applications, especially where high impedance, capacitive transducers are used The high input impedance JFET front end makes the such as hydrophones, precision accelerometers, and LT1169 suitable in applications where very high charge photodiodes. The total output noise in such a system is sensitivity is required. Figure 3 illustrates the LT1169 in its the gain times the RMS sum of the op amp’s input referred inverting and noninverting modes of operation. A charge voltage noise, the thermal noise of the transducer, and the amplifier is shown in the inverting mode example; the gain op amp’s input bias current noise times the transducer depends on the principal of charge conservation at the impedance. Figure 2 shows total input voltage noise input of the LT1169. The charge across the transducer versus source resistance. In a low source resistance capacitance CS is transferred to the feedback capacitor CF (< 5k) application the op amp voltage noise will dominate resulting in a change in voltage dV, which is equal to dQ/CF. The gain therefore is 1 + CF/CS. For unity-gain, the CF 10k C should equal the transducer capacitance plus the input S LT1124* capacitance of the LT1169 and RF should equal RS. √Hz) – RS LT1169* 1k + In the noninverting mode example, the transducer current VO is converted to a change in voltage by the transducer R C S S 100 capacitance, CS. This voltage is then buffered by the LT1124† LT1169 with a gain of 1 + R1/R2. A DC path is provided by LT1169 R 10 S, which is either the transducer impedance or an LT1169† INPUT NOISE VOLTAGE (nV/ external resistor. Since RS is usually several orders of LT1124 magnitude greater than the parallel combination of R1 RESISTOR NOISE ONLY 1 100 1k 10k 100k 1M 10M 100M 1G and R2, RB is added to balance the DC offset caused by the SOURCE RESISTANCE (Ω) noninverting input bias current and RS. The input bias LT1169 • F02 currents, although small at room temperature, can create SOURCE RESISTANCE = 2RS = R * PLUS RESISTOR significant errors over increasing temperature, especially † PLUS RESISTOR  1000pF CAPACITOR with transducer resistances of up to 1000MΩ or more. V 2 n = AV √Vn (OP AMP) + 4kTR + 2qIBR2 The optimum value for RB is determined by equating the
Figure 2. Comparison of LT1169 and LT1124 Total Output
thermal noise (4kTRS) to the current noise (2qIB) times
1kHz Voltage Noise vs Source Resistance
R 2 S . Solving for RS results in RB = RS = 2VT/IB. A parallel 8